Method and apparatus for distortion correction of RF amplifiers

ABSTRACT

A method of reducing distortion in the output of an amplifier is provided. The method comprises subtractively combining an error signals with the appropriate phase shift with input signals to be amplified. The error signal being generated by subtractively combining a fed-forward portion of the input signal with a portion of the fed-back amplified output signal, and signal processing applied to it between its generation and application to correcting the input signal in the baseband domain. The error therefore being down-converted, filtered, and up-converted in the feedback path. The filtered baseband error signal components providing inputs to a controller which adjusts active elements of the amplification and feedback path in order to minimize the distortion within the output of the amplifier.

FIELD OF THE INVENTION

The invention relates to microwave amplifiers, and more specifically tocorrecting distortion generated in microwave amplifiers.

BACKGROUND OF THE INVENTION

In recent years, advances in semiconductor integrated circuits forwireless and RF technology have dramatically changed our perceptions,use, and reliance upon portable electronic devices. The uses of wirelesstechnology are widespread, increasing, and include, but are not limitedto, telephony, Internet e-mail, Internet web browsers, globalpositioning, photography, diary, address book, and in-store navigation.Additionally, devices incorporating wireless technology have expanded toinclude not only cellular telephones, but Personal Digital Assistant(PDAs), laptop computers, palmtop computers, gaming consoles, printers,telephone headsets, portable music players, point of sale terminals,global positioning systems, inventory control systems, and even vendingmachines. Today, many of these devices are high volume consumercommodities supplied by businesses competing through integrated featuresand rapidly changing branding whilst reinforcing customers desire forsmall size, long battery life, and lightweight devices with increasedroaming capabilities and download speeds. Within a matter of a fewyears, these systems have evolved from bulky cellular telephonesoffering voice and simple text messaging to lightweight compactmulti-media players providing streaming live video and music alongsidetelephony, PDA functionality, integrated mega pixel CCD camera, andsupporting BlueTooth™ wireless peripheral interfaces for headphones,microphones etc.

Semiconductor integrated circuits have been an important enabler of thisrapid evolution by offering high volume, low cost solutions with lowpower consumption, small footprint and reduced component count whencompared to discrete or hybrid solutions. There is significant economicand business pressure to ensure that these trends continue, whilstproviding increased benefits to the manufacturers including reducing thenumber of chips for a chip set, providing multiple standards from asingle chip set, and providing specifications and margin allowingtoday's chip sets to support the evolving bandwidth and spectral aspectsof these systems and standards.

Indeed, such wireless semiconductor circuits today address a plethora ofstandards including, but not limited to, IEEE 802.11 WiFi, IEEE 802.16WiMAX, quad-band GSM, EDGE, GPRS, and Global Positioning Systems. Inmany instances, such as IEEE 802.16e WiMAX with targeted data rates of10 Mb/s at a 10 km range from a base station, the systems are alsostretching the limits of performance in respect of transmitted power,received power, dynamic range, interference, efficiency and bandwidth.

All of these performance aspects impact the design and implementation ofthe RF amplifiers that form a critical element in both the transmitterand receiver portions of the transceivers within these devices. In orderto increase efficiency wireless amplifiers have tended to move from thehigh linearity amplifier design typified by classes A, AB, and B,(http://en.wikipedia.org/wiki/amplifier) to non-linear amplifier classessuch as C, D, E and F, where efficiencies over 90% can be achieved.

The evolution from low efficiency, high linearity amplifiers to veryhigh efficiency non-linear amplifiers has circuit designers exploitingclassical techniques to correct the distortion generated in theamplifier. For those skilled in the art such classical techniques fallinto 5 different classes; feed-forward, feedback, pre-distortion,adaptive bias, and synthesis. However, the basic objective on anyamplifier in the context of a wireless information transmission systemis to provide an exact copy of the signal intended to be transmitted atthe correct power level with highest possible power efficiency.

Feed-Forward: This approach employs an error signal that is extractedfrom an amplifier, commonly referred to as the power amplifier (PA) onthe transmit path. Considering a PA for the following discussions, thenthe PA is corrected by subtracting a scaled (attenuated) version of theoutput signal of the PA from the input signal. If these signals areproperly scaled, then the resulting signal contains only errorinformation that is spectral energy generated by the non-linearity ofthe PA, noise from the PA, and energy resulting from non-flat frequencyresponse of the PA, and none of the original signal. This error signalis then amplified appropriately, and subtracted from the output signalof the PA. Typically, the output signal from the amplifier is timedelayed to account for the delay in the components of the error signalgeneration path. An example of the feed-forward approach is disclosed byChen et al in “Article comprising a Power Amplifier with Feed ForwardLinearizer using a RLS Parameter Tracking Algorithm” (U.S. Pat. No.5,963,091) and described in respect of FIG. 1 subsequently. If theamplitude and phase are correct, feeding forward and subtracting theerror removes all the error (distortion) generated in the PA in questionand is powerful in that the approach corrects any error generated by thePA. However, it is a very inefficient correction technique as firstly,the amplifier boosting the error signal must itself be fairly large andlinear as its output signal is generally combined with the PA outputsignal using a low ratio coupler to avoid losses on the main transmitpath. Secondly, the time delay applied to the output signal of the PAdue to the circuit delay in the error signal path can add significantloss.

Pre-Distortion: This correction approach employs a model of the PA topredict the distortion that will be generated in the amplifier beingcorrected. The modeled distortion is then added to the input signalprovided to the amplifier, with appropriate phase adjustment, such thatit cancels the distortion produced. The model can take many formsincluding, but not limited to, a look-up table of amplitude modulation(AM) and phase modulation (PM) transfer curves such as AM-AM and AM-PMcurves, it can be non-linear electronic hardware, it can be DSPalgorithms, or it can be a scaled model of the amplifier beingcorrected. An example of pre-distortion applied to an amplifier ispresented by Midya et al “Scalar Cost Function based Pre-DistortionDevice, Method, Phone and Base Station” (U.S. Pat. No. 6,240,278), aspresented and discussed in respect of FIG. 2.

Pre-distortion is an efficient technique, in that there are no lossycircuit elements after the power amplifier, and there are no additionalmicrowave circuit blocks that are inherently power-hungry. Further, theability to use digital hardware, which has dramatically improved incapability in recent years, has made this a favored solution. However,predistortion can only easily cope with simple memoryless deterministicdistortion and typically assumes that the AM-AM and AM-PM curves arestatic and do not depend on earlier events, operational conditions, orfrequency of operation. Furthermore, the nature of the model used topredict the error must suit the amplifier. That is, the model mustemploy an appropriate order of non-linearity, or an appropriate numberof entries in the look-up table.

This makes the pre-distortion system somewhat specific to the amplifierbeing corrected. Although digital hardware tends to be relatively lowcost, a digital pre-distorter can be quite elaborate, requiring widedata bus and fast sample rates (usually at least 5× the Nyquist rate ofthe data rate within the uncorrected signal). Finally, the algorithmsfor adjusting the non-linearity (the weights on the non-linear workfunction or the elements in the look-up table) are complex and prone tofinding local minima. Despite these limitations and disadvantages,pre-distortion has a significant share of the distortion correctionsolutions implemented today.

Adaptive Bias: In contrast to the previous approaches, adaptive biastechnique does not attempt to correct distortion products but rather canimprove either the linearity or minimize the distortion of the PA. In anadaptive bias system, the bias voltages on the terminals of the activedevice are adjusted to suit the instantaneous signal being transmitted.For example, the collector or drain voltage of a transistor amplifiercan be increased during peaks in the amplitude of the input signal. Thistechnique can be a simple way to make modest improvements to amplifierlinearity, however it also is amplifier dependant, and largeimprovements in linearity are difficult to achieve. As a result, theadaptive bias approach has limited benefit to the very high efficiencybut highly non-linear amplifier classes which suit the demands for lowpower consumption in wireless handheld devices.

Synthesis: This approach is more a general category of “synthesistechniques”, in which a linear PA is not used, but instead the signal atthe output port of the amplification system is generated by combining 2or more signals, each of the initial signal components not resemblingthe final signal being generated. Examples of this technique includeEnvelope Elimination and Restoration (EE&R), as discussed by Midya, Khanet al “Method, Device, Phone and Base Station for ProvidingEnvelope-Following for Variable Envelope Radio Frequency Signals” (U.S.Pat. No. 6,141,541), and Linear Amplification with Nonlinear Components(LINC), as discussed by Okubo et al “Constant-Amplitude Wave CombinationType Amplifier” (U.S. Pat. No. 5,287,069).

In EE&R, a constant amplitude signal with variable phase is amplified,and the envelope restored by varying the collector or drain voltage ofthe transistor amplifier. In LINC, two constant amplitude signals arecombined in various phases to generate a signal with the correctamplitude and phase. These techniques have specific applications, butall have significant drawbacks in respect of power, bandwidth,efficiency and linearity.

Feedback: The general technique of feedback correction goes back nearly80 years, see for example H. S. Black “Wave Translation System” (U.S.Pat. No. 2,102,671; filed 1932). In particular, negative feedback tendsto act to reduce variability of gain, and reduces distortion introducedin an amplifier. The actual feedback may be implemented in manydifferent forms. Perhaps the simplest form of feedback being referred toas circuit-level feedback, wherein an electrical linkage couples some ofthe energy from the output port of an amplifier back to the input port.Considering a single transistor amplifier, such approaches includeshunt-feedback, where a resistor is placed between the drain and gate ofa transistor, and series feedback, where an inductor is inserted intothe source of a transistor.

Circuit level feedback can be applied to multiple stages, but the signaldelay through the stages must be accounted for. If there is too muchdelay, then the negative feedback will, at some frequency, becomepositive feedback, and an oscillation results because a portion of theoutput signal (at a particular set of frequencies) adds constructivelyto the input signal. With each passage through the loop, the signalincreases to the point where all energy can be found in those particularfrequencies where the combination is most constructive. Of course, aslong as the gain is less than unity at the frequency at which the phaseshift through the feedback loop is 360 degrees, the amplifier willremain stable. While simple circuit level feedback is widely used itsuffers from one major fault in that it necessarily decreases the gainof the amplifier. Hence, higher levels of correction necessarilyincreasing gain which is an issue at RF frequencies where gain isdifficult and expensive to achieve.

As a result, other feedback approaches have been established to act upononly the information on the envelope of the signal being amplified,which has the advantages that the RF gain of the amplifier isundiminished, and the feedback circuit may be implemented in baseband.Two such approaches being Cartesian Feedback and Polar Feedback.Considering, Cartesian Feedback, a typical approach is presented byLeitch “Gain/Phase Compensation for Linear Amplified Feedback Loop”(U.S. Pat. No. 4,933,986), in demodulating the signal at the output portof the amplifier into in-phase (I) and out-of-phase (Q) components inthe Cartesian coordinate system. These I- and Q-output signal componentscan be compared to the I-and Q-input signals, and the results applied toan I/Q modulator at the input port of the amplifier to providecorrection.

An advantage of the Cartesian feedback approach is that the filteringthat keeps the amplifier stable can now be done at baseband, which isadvantageous, as it no longer needs to be tuned with the RF frequency ofoperation of the amplifier. The design of this filter is critical,however, as the filter's frequency response is superimposed onto thegain of the amplifier. Further, delay through the amplifier is critical,as it determines the bandwidth of the filter required to stabilize thefeedback loop, and therefore the instantaneous bandwidth of the system.

In contrast, Polar feedback resolves the amplitude and phase elements ofthe output signal, compares these to the amplitude and phasecharacteristics of the input signal, and uses the resulting error termsto control a polar modulator placed before the input port of the poweramplifier, thereby closing the loop. This approach is presentedsubsequently in FIG. 4. Both Cartesian and Polar feedback techniquessuffer from the limitations of the loop filter that is used to ensurethat the gain of the loop rolls off fast enough with increasingfrequency such that the loop remains stable. In the prior art, this loopfilter adds delay and amplitude variation with frequency across theband.

It is possible, however, to configure feedback loops in which just theerror is fed back, such an approach being presented by Huang “Wideband,Phase Compensated Amplifier with Negative Feedback of DistortionComponents in the Output Signal” (U.S. Pat. No. 4,276,514). Thisapproach generates the error term by subtracting the input signal fromthe attenuated version of the output signal. This error term is fed backinto the input port of the amplifier in anti-phase (inverted) to effectthe correction. This is advantageous, as the frequency response of thefilter is now interposed only on the error signal and the main signal isnot at all affected. Ripple in the frequency response of this filterwill vary the amount of correction applied, but not the gain of the mainsignal. Implementations of error feedback have been limited, primarilyas initial publications on error feedback, such as John McRory “An RFAmplifier for Low Intermodulation Distortion” (1994 MTT-S Digest, pp.1741-) resulted in minimal improvements being observed. Two significantchallenges in implementing this architecture exist. Firstly, theimplementation of the tunable filter which must be at RF, but have afinesse adequately high to maintain stability, and secondly in designingthe components in the loop to have minimal delay so that a usefulcorrection bandwidth results.

The classical techniques presented supra all suffer from one or more ofthe following impairments:

-   -   Poor Efficiency    -   Limited Bandwidth    -   Complexity    -   Limited Effectiveness

It would be apparent that such limitations result in system designersand circuit designers trading aspects of performance and cost inimplementing commercial systems with non-linear amplifiers for improvedefficiency using these prior art approaches.

It would therefore be advantageous to provide a linearization solutionfor an RF amplifier that addresses these drawbacks of prior artapproaches whilst leveraging increased integration potential withinsemiconductor integrated circuits for lowering cost, footprint and powerconsumption. It would be particularly beneficial if the linearizationsolution addressed the increasing fractional bandwidth of today'sincreasing data rate wireless protocols.

SUMMARY OF THE INVENTION

In accordance with the invention there is provided a method comprisingamplifying an applied RF input signal to generate an RF output signalwith an amplifier, generating an error signal by combining apredetermined portion of the RF output signal with a predeterminedportion of a supplied RF input signal, and modifying the error signal togenerate a modified error signal, the modified error signal generated bydown-converting the error signal, processing the down-converted errorsignal and up-converting the processed down-converted error signal. Themethod further comprising reducing distortion in the RF output signal bycombining the modified error signal with the supplied RF input signal toprovide the applied RF input signal, and controlling an aspect of atleast one of the generation of the modified error signal, generation ofthe error signal, and the applied RF input signal at least in dependenceupon a magnitude of the down-converted error signal.

In accordance with another exemplary embodiment of the invention thereis provided a circuit comprising an amplifier for amplifying an appliedRF input signal to generate an RF output signal, and an error signalcircuit for generating an error signal by combining a predeterminedportion of the RF output signal with a predetermined portion of asupplied RF input signal. The circuit further comprises an error signalmodification circuit for modifying the error signal to generate amodified error signal, the modified error signal generated bydown-converting the error signal, processing the down-converted errorsignal and up-converting the processed down-converted error signal, andan error combiner circuit for reducing distortion in the RF outputsignal by combining the modified error signal with the supplied RF inputsignal to provide the applied RF input signal. The circuit furthercomprises a controller for controlling an aspect of at least one of thegeneration of the modified error signal, generation of the error signal,and the applied RF input signal at least in dependence upon a magnitudeof the down-converted error signal.

In accordance with another exemplary embodiment of the invention thereis provided a method comprising the steps of amplifying a microwavesignal with at least an amplifier to provide an output signal, theoutput signal being substantially an amplified replica of the microwavesignal, and the step of providing an input signal, the input signalforming a portion of the microwave signal to be amplified. The methodfurther comprising the steps of extracting a predetermined portion ofthe output signal, and the step of combining a predetermined portion ofthe input signal with the predetermined portion of the output signal toform an error signal, the error signal substantially suppressing theinput signal component and substantially unaffecting distortioncomponents of the output signal.

The method further comprises the step of down-converting the errorsignal according to a predetermined coordinate system to providedown-converted error components, and the step of processing thedown-converted error components to provide modified down-converted errorcomponents, and the step of up-converting the modified down-convertederror components according to the predetermined coordinate system toprovide a modified error signal. Further the method comprises the stepof combining the modified error signal with the input signal to form themicrowave signal and provide a reduction in the distortion components ofthe output signal, and the step of providing at least a control signal,the control signal being one of a plurality of control signals, eachcontrol signal for controlling an aspect of at least one of steps (a)through (h) and being determined in dependence upon a measure of one ofthe down-converted error components and modified down-converted errorcomponents.

In accordance with another exemplary embodiment of the invention themethod further comprises providing an electrical circuit, and providinga first predetermined portion of the electrical circuit as portion of afirst integrated circuit, and providing a second predetermined portionof the electrical circuit as a portion of a second integrated circuit.The method further comprises the step of assembling the first and secondintegrated circuits, the assembly minimizing the time delay from theamplifier output port to the amplifier input port.

In accordance with another exemplary embodiment of the invention thereis provided a method further comprises providing an electrical circuit,providing a first predetermined portion of the electrical circuit as aportion of a first circuit package, and providing a second predeterminedportion of the electrical circuit as a portion of a second circuitpackage. The method further comprising assembling the first and secondcircuit packages, the assembly reducing the time delay from theamplifier output port to the amplifier input port.

In accordance with another exemplary embodiment of the invention themethod comprises providing predetermined portions of an electricalcircuit implementing the method as portions of circuit packages, whereinproviding the portions of the circuit packages comprises placing eachrespective portion of the electrical circuit within a circuit packagethereby reducing the time delay associated with that portion of theelectrical circuit and its respective circuit package.

BRIEF DESCRIPTION OF THE DRAWINGS

Exemplary embodiments of the invention will now be described inconjunction with the following drawings, in which:

FIG. 1 illustrates an exemplary feed-forward distortion correctionapproach according to the prior art.

FIG. 2 illustrates an exemplary DSP based pre-distortion correctionapproach according to the prior art.

FIG. 3 illustrates an exemplary RF error feedback correction approachaccording to the prior art.

FIG. 4 illustrates an exemplary polar loop feedback distortioncorrection approach according to the prior art.

FIG. 5 illustrates a first exemplary embodiment of the inventionproviding distortion correction by RF feedback and baseband I and Qsignal processing of the error signal.

FIG. 6 illustrates a second exemplary embodiment of the inventionproviding distortion correction by RF feedback and baseband I and Qsignal processing of the error signal operating over wide dynamic rangeand continuous control.

DETAILED DESCRIPTION OF EMBODIMENTS OF THE INVENTION

Shown in FIG. 1 is an exemplary feed-forward amplifier system 100according to Chen et al (U.S. Pat. No. 5,963,091). As shown, thefeed-forward distortion correction system 100 comprises an RF input port100A which receives an RF modulated signal Vm which is to be amplified,and provided as an amplified linear replica at the output port 100B asamplified signal Vo.

The RF input port 100A is electrically coupled to the input splitter 105which taps a predetermined portion of the RF modulated signal Vm to feedforward into the correction signal path 1100, and couples the remainingRF modulated signal Vm into the main arm 1200. The main arm 1200 couplesthe RF modulated signal Vm from the input splitter 105 to the poweramplifier 101 which amplifies the RF modulated signal Vm as required bythe overall system. The feed-forward amplifier system 100 providesamplification, that is required, for example in the transmit path of awireless WiMAX transceiver embedded into a portable device. The outputsignal from the power amplifier 101 is coupled to a signal splitter 102wherein a predetermined portion of the amplified signal is tapped andcoupled to the correction signal path 1100 by virtue of beingelectrically coupled to the attenuator 107. The remaining output portfrom the signal splitter 102 is then electrically coupled to a firstdelay element 103 and thence to cancellation adder 104 before beingelectrically connected to the RF output port 100B.

The portion of the RF modulated signal Vm split from the input splitter105 and provided to the correction signal path 1100 is initially coupledto a second delay element 106 before being coupled the first port of theadder circuit 109. The second port of the adder circuit 109 is coupledto the cancellation vector modulator 108 that receives the attenuatedportion of the amplified RF signal from the attenuator 107. In thismanner, the adder circuit 109 receives a time-delayed replica of the RFmodulation signal Vm and a replica of the amplified signal from thepower amplifier 101. As such, with appropriate control of the attenuator107 and cancellation vector modulator 108, the portions of each replicasignal being of equal magnitude and out of phase such that the addercircuit 109 provides an output signal which is solely the errorintroduced by the power amplifier 101, Vd.

This error signal Vd is then electrically coupled to a cancellationerror vector modulator 110 and auxiliary amplifier 111, the output portof the auxiliary amplifier 111 being coupled to the other input port ofthe cancellation adder 104. In a similar manner to the attenuator 107and cancellation vector modulator 108, the intent with the cancellationerror vector modulator 110 and auxiliary amplifier 111 is provide anerror signal of equal magnitude to the error signal components withinthe main arm 1200 from the power amplifier 101, and one phase shiftedsuch that when added to the amplified RF modulated signal within thecancellation adder 104 the error signal is cancelled, resulting in anoutput RF signal Vo which is a highly linear, amplified replica of theRF modulation signal Vm.

A digital signal processor 115 is configured to receive input signalsdetermined in dependence upon the RF modulation signal Vm, output RFsignal Vo and error signal Vd at its control input ports 115 a through115 c. These are down-converted using down-converter circuit 116, whichis configured to shift the frequency range of the signals, Vm, Vd, andVo into the baseband frequency range. These down-converted signals arethen coupled to the digital signal processor (DSP) 113, which performsthe necessary calculations to determine appropriate control signals toelements of the correction signal path 1100. The DSP 113 is also incommunication with memory device 114 which provides data to the DSP 113.The DSP 113 provides data to the D-A converter 112, which has two DACoutput ports 112 a and 112 b. The D-A converter 112 converts calculatedparameters for the correction factors to first analog signal α that iscoupled to the cancellation vector modulator 108 from DAC output port112 a, and second analog signal β that is coupled to the cancellationerror vector modulator 110. The DSP 113 thereby monitors the RF outputsignal Vo, and determines adaptions to the first and second analogsignals α and β.

Feed-forward correction is a powerful technique, in that it corrects anyerror generated by the power amplifier 101. However, the correctiongenerated and fed forward by correction signal path 1100 must beamplified to the same power level as that of the power amplifier 101. Asa result, the auxiliary amplifier 111 is typically a fairly largeamplifier due to the typically low split ratio of the cancellation adder104. This results in high power consumption. Additionally, the auxiliaryamplifier must be an amplifier of high linearity as it is amplifying thecorrection signal, and is itself uncorrected.

Referring to FIG. 2 shown is an exemplary DSP pre-distortion amplifiercircuit 200 according to Midya et al (U.S. Pat. No. 6,240,278).Accordingly shown is an input port 200A which receives a baseband signalto be amplified (I+jQ) which is electrically connected to a polynomialpre-distortion unit 201, the output signal of which is a predistortedbaseband signal (I′+jQ′) applied to the input port of a RF modulator 202which translates the predistorted baseband signal to an RF input signaland couples this to the input port of the power amplifier 203. Theoutput port of the power amplifier 203 is electrically coupled to theoutput port 200B via a splitter 204. The power amplifier 203 is biasedfrom the power supply 208.

The tapped portion of the amplified RF signal from the splitter 204 iselectrically coupled to a scalar function generator 206, which computesa scalar out-of-band-energy function and provides this to thecoefficient update unit 205. The scalar function generator 206comprising at least an RF mixer, not shown for clarity, to convert theRF output signal to baseband for digital processing. The coefficientupdate unit 205 determines whether any tuning of the polynomialcoefficients applied by the polynomial pre-distortion unit 201 isrequired, the coefficients being determined to minimize the scalarout-of-band-energy function to provide a linear amplified RF signal.

As taught by Midya et al the polynomial pre-distortion unit 201pre-distorts the applied baseband signal using the third and fifth orderterms of the non-linear component of the output signal from theamplifier, referred to as p3 and p5 respectively, which are the productsof the complex baseband signal and the predetermined scalar function ofthe power transfer characteristic of the power amplifier 203. Theseterms are shown below in Eq. 1 and Eq. 2 respectively.

p3(I,Q)=(I+jQ){(I ² +Q ²)}; and   [1]

p5(I,Q)=(I+jQ){(I ² +Q ²)²}  [2]

These predetermined terms of the power series are weighted to adjustaccording to the given power amplifier 203, the power series coefficientweights (w_(k)) being complex. This accounts for the amplitudemodulation (AM) and phase modulation (PM) nonlinearity as well as theAM-AM nonlinearity correction from the real part of the coefficients.Thus the predistortion may have the format presented below in Eq. 3.

(I′+Q′)=(I+jQ)+w ₃ p ₃ +w ₅ p ₅   [3]

It is these power series coefficient weights (w_(k)) as a series ofpolynomial coefficients that are provided from the coefficient updateunit 205 to the polynomial pre-distortion unit 201. It would be evidenttherefore that the polynomial pre-distortion approach requires detailedevaluation of the power amplifier 203 and derivation of the appropriatepolynomial series for the correction. Hence, whilst the approach employssolely baseband digital processing, the approach does not easily accountfor aging effects within the power amplifier that would render theexisting coefficients less than optimal in reducing distortion. Whilstthe power series coefficient weights (w_(k)) may be varied limits orflexibility in setting them may not correct for all aging effects.Further, variations in the voltage provided from the power supply 208,either intentionally to provide gain control, or unintentionally canresult in the need for the digital circuits storing and employingmultiple power series.

Since typically predistortion circuits such as DSP pre-distortionamplifier circuit 200 do not have lossy elements after the poweramplifier 203, with the exception of the splitter 204, the approach ispower efficient. However, the DSP pre-distortion amplifier circuit 200approach disclosed by Midya et al is atypical amongst predistortioncircuit prior art in that it comprises a feed-back element of thepredistortion circuitry. As such, the DSP pre-distortion amplifiercircuit 200 can provide some limited adjustment in correction appliedwith variations in the power amplifier 203 performance, these being thepower series coefficient weights, w_(k). However, such techniques canstill only cope with simple memoryless deterministic distortion, employmodels that assume amplification distortion curves are static, and mustemploy models with either appropriate order of nonlinearity or number ofentries within the loop-up table to provide correction over the range ofoperation of the power amplifier 203.

Alternative to performing feed forward or pre-distortion many havesought to provide distortion correction by directly feeding back theoutput signal to the input port of the power amplifier. Such an approachbeing shown in FIG. 3 wherein an RF signal to be amplified, RF_(in), isapplied to the input port 300A of an RF feedback circuit 300, and iselectrically connected to the input splitter 310. The primary outputport 310A of the input splitter is electrically connected to first port320A of the summation circuit 320. The secondary output port 310B of theinput splitter 310 is connected to the first port 380A of the deltagenerator circuit 380.

The output port of the summation circuit 320 is connected to the inputport of the power amplifier 330, wherein it is amplified, andelectrically coupled to the output port 300B of the RF feedback circuit300 via the output tap coupler 340. The second output port of the outputtap coupler 340 represents a predetermined portion of the output signalfrom the power amplifier 330 and is therefore an attenuated replica ofthe output signal, RF_(out), at the output port 300B. The output signalbeing RF_(out)=

(RF_(in))+∂, where

( ) represents the amplification process and ∂ represents distortionadded by the power amplifier 330.

The second port of the output tap coupler 340 is then coupled to anattenuator 390 and electrically coupled to the second port 380B of thedelta generator circuit 380. The attenuator 390 thereby further reducingthe power of the replica of the output signal, RF_(out), toapproximately that of the tapped portion of the RF signal, RF_(in).Accordingly the delta generator circuit 390 produces an output signal atoutput port 380C which essentially represents distortion generatedwithin the power amplifier 330 as the input signal, RF_(in), issubtracted from the tapped and attenuated replica of the output signal,RF_(out).

This output signal ∂ ∝

RF_(out)−RF_(in)

is then electrically coupled via the feedback arm 3100 to the secondport 320B of the summation circuit 320. As a result the output signalfrom the summation circuit 320 is SUM=RF_(in)−∂, such that the poweramplifier 330 input signal is corrected for the distortion ∂ addedduring the amplification process

( ). The feedback arm 3100 comprises a feedback amplifier 370, whichboosts the output signal ∂ to a level appropriate for summation with theinput signal in the summation circuit 320, after which it is filtered bylow pass filter 360 and phase shifted within the phase shifter circuit350. The low pass filter 360 provides stability to the feedback circuitby reducing the gain of frequencies at which the phase around thefeedback loop reaches zero, to below unity such that oscillation issuppressed. The phase shifter circuit 350 providing any additional phaseshift to the feedback signal, the distortion ∂, such that within theoperating frequency range of interest the signal applied to thesummation node 320 is of the correct phase to cancel the distortiongenerated in the amplifier.

The RF feedback circuit 300 of FIG. 3 generates, manipulates and appliesthe RF feedback signal directly in the microwave domain, therebyproviding a simple circuit implementation. An alternative approach,presented in FIG. 4 applies the distortion signal ∂ via modulation ofthe input signal, and within this example uses polar modulation toaccomplish this. As shown, an RF signal RF_(in), is applied to the inputport 400A of a polar loop feedback circuit 400, and is electricallyconnected to the input splitter 410. The primary output port 410A of theinput splitter 410 is electrically connected to the main signal arm4100, and the tap output port 410B of the input splitter 410 iselectrically connected to the polar loop feedback arm 4200, and providesthe first signal into the polar loop feedback arm 4200. The first signalelectrically coupled from the tap output port 410B is electricallyconnected to the first diode detector 470 and phase comparator 480. Themain signal arm 4100 comprises variable attenuator 420 and variablephase modulator 425 electrically connected in series between the primaryoutput port 410A and the power amplifier 430. The output port of thepower amplifier 430 is then electrically connected to the outputsplitter 415. The output splitter 415 is electrically coupled to theoutput port 400B, thereby providing the amplified RF output signal,RF_(out). The second port of the output splitter 415 provides the secondsignal into the polar loop feedback arm 4200.

The second port of the output splitter 415 is electrically connected tothe feedback attenuator 460, the output port of which is electricallyconnected to the second diode detector 475 and phase comparator 480. Theoutput signals from the first and second diode detectors 470 and 475 areelectrically connected to the positive and negative input ports ofdifferential amplifier 450, the output port of which is electricallycoupled to the control port 420A of the variable attenuator via thefirst loop filter 440. As the first diode detector 470 detects thereceived RF signal RF_(in), and the second diode detector 475 detectsthe amplified RF output signal, RF_(out), the output signal from thedifferential amplifier 450 is directly determined by the errorintroduced by the power amplifier 430.

The output port from the phase comparator 480 is electrically connectedto the control amplifier 455, and thence to the control port 425A of thevariable phase modulator 425 via the second loop filter 445. As aresult, the phase comparator 480 and differential amplifier 450, inconjunction with first and second diode detectors 470 and 475, providefor independent control of the amplitude and phase of the RF signal,RF_(in), immediately prior to the power amplifier 430. In this manner,the polar loop feedback arm 4200 provides correction of the distortionintroduced by power amplifier 430.

According to the correction circuits presented supra in FIGS. 3 and 4the bandwidth of the feedback correction circuits is determinedsubstantially upon the delay within the feedback loop. As such, signalphase changes rapidly with frequency such that it is necessary to reducethe gain of the loop at frequencies wherein the phase shift would causethe loop to oscillate. As requirements for wider bandwidths arising fromnew systems become standard and evolve, such correction approaches willtypically require a tradeoff in correction performance with bandwidth.Referring to FIG. 5 shown is an exemplary first embodiment of theinvention wherein this tradeoff is modified such that wider bandwidthscan be corrected whilst preserving the gain of the RF signal paththrough the amplifier.

As shown in FIG. 5, an RF signal to be amplified, RF_(in), is applied tothe input port 500A of an RF feedback circuit 500, and is electricallyconnected to the input splitter 510. The primary output port 510A of theinput splitter 510 is electrically connected to first port 520A of thesummation circuit 520. The secondary output port 510B of the inputsplitter 510 is connected to the first port 545A of the delta generatorcircuit 545.

The output port of the summation circuit 520 is connected to the inputport of the power amplifier 530 via the first phase shifter circuit 525.The RF input signal, RF_(in), is amplified by the power amplifier 530,and electrically coupled to the output port 500C of the RF feedbackcircuit 500 via the output tap coupler 515. The second output signal ofthe output tap coupler 515 represents a predetermined portion of theoutput signal from the power amplifier 530 and is therefore anattenuated replica of the output signal, RF_(out), at the output port500C. The output signal being RF_(out)=

(RF_(in))+∂_(err), where

( ) represents the amplification process and ∂_(err) representsdistortion added by the power amplifier 530.

The second output port of the output tap coupler 515 is then coupled toan attenuator 540 and electrically coupled to the second port 545B ofthe delta generator circuit 545. The attenuator 540 thereby furtherreduces the power of the replica of the output signal, RF_(out), toapproximately that of the tapped portion of the RF signal, RF_(in).Accordingly the delta generator circuit 545 produces an output signal atoutput port 545C which essentially represents distortion generatedwithin the power amplifier 530 as the input signal, RF_(in), issubtracted from the tapped and attenuated replica of the output signal,RF_(out).

This output signal ∂≈∂_(err)∝<RF_(out)−RF_(in)

is then electrically coupled to the feedback processing circuit 5100.The output signal ∂ is first coupled to the vector demodulator 560 thatconverts this output signal ∂ to I and Q baseband error signals ∂_(I)and ∂_(Q), which are output signals from 1 error port 560A and Q errorport 560B, respectively. The I baseband error signal ∂_(I) iselectrically coupled via first baseband amplifier 572 before beingfiltered by first baseband filter 575 and coupled to the I input port590A of the vector modulator 590. Similarly, the Q baseband error signal∂_(Q) is electrically coupled via second baseband amplifier 570 beforebeing filtered by second baseband filter 574 and coupled to the Q inputport 590B of the vector modulator 590. The output port of the vectormodulator 590 is boosted by the error amplifier 595 before beingelectrically coupled to the second port 520B of the summation circuit520. In this manner, the error signal ∂ generated by the power amplifier530 is combined with the correct phase relationship to the RF signal tobe amplified RF_(in) to provide correction for operation of the poweramplifier 530.

The local oscillator signal required by the vector demodulator 560 andvector modulator 590 is provided by the local oscillator 585. Whilst thelocal oscillator 585 is shown directly coupled to the vector modulator590 it is electrically connected to the vector demodulator 560 via asecond phase shifter circuit 580, which ensures the appropriate phase ofthe local oscillator signal to demodulate the I and Q signals ∂_(I) and∂_(Q) respectively.

Fulfilling high loop gain, for high distortion suppression, whilstavoiding oscillation is challenging, as will be evident to one skilledin the art, because the amplifier may have several stages, and narrowband matching, so that the phase through the amplifier changes quicklywith frequency, in other words, the amplifier may have significantdelay. As such, stability of the feedback loop comprising substantiallyof the feedback processing circuit 5100 is provided by the first andsecond baseband filters 574 and 575 respectively, these reducing theoverall gain of the feedback loop to below unity for frequencies whereinthe phase shift around the feedback processing circuit 5100 and elementsof the amplification path between the summation circuit 520 and deltagenerator circuit 545 is approximately zero. However, unlike prior artsolutions these first and second baseband filters 574 and 575,respectively are only filtering the baseband error signal generatedwithin the feedback processing circuit 5100. Beneficially, the criticaldesign aspect of this filter is transformed from ‘flatness of frequencyresponse’ to the provisioning of sharp cut-off allowing high in-bandloop gain but rapid reduction in out-of-band gain as required to meetloop stability requirements.

Additionally, to successfully configure this amplifier, it is necessaryto minimize the total loop delay so that the first and second basebandfilters 574 and 575 can be provided with wide bandwidth since a wideloop filter allows correction over a wide bandwidth of signal, andmaintain the loop gain sufficiently large enough to effect mid-bandcorrection. This is particularly important with WiMAX and WiFiapplications where channel bandwidths of 10 MHz and 20 MHz,respectively, are desired, such wide bandwidths thereby limiting overallloop delay. Benefit provided by the Cartesian error feedback circuit 500increases according to the design attention in respect of minimizingthis loop delay.

Whilst the exemplary embodiment of the invention has been presented withrespect to a Cartesian error feedback method, alternatively othercoordinate systems for the provision of baseband signals may beemployed. As such, the error feedback approach of the invention may beapplied to polar coordinate modulation formats, and optionally arbitrarycoordinate modulation formats.

With WiMAX subscriber units having over 50 dB of dynamic range in outputpower, the PA within such units will typically operate from very lowgain and output power, wherein the main power amplifier will typicallybe relatively linear, to high gain and high output power, wherein itwill be typically highly non-linear. As such, there is benefit inoperating the RF feedback circuit 500 in two or more modes of operationaccording to output power. Optionally, these modes may be switchedaccording to other aspects of the system within which the RF feedbackcircuit 500 is embedded, such as managing power consumption in handhelddevices. As such, at high output powers, with highly non-linearamplifier characteristics the RF feedback circuit 500 operates aspresented in respect of the embodiment presented in respect of FIG. 5.At low output powers, where the power amplifier 530 has typicallysignificantly improved linearity the feedback processor circuit 5100along with the delta generator circuit 545, local oscillator 585 andsecond phase shift circuit 580 can be disabled by removing appliedelectrical power or by other means known in the art, such that the RFfeedback circuit 500 now operates in an open-loop but with reduced powerconsumption. In instances where switching between modes is in a timethat is shorter than the time that the local oscillator 585 can bestabilized it may be optionally left on in both modes.

Referring to FIG. 6, illustrated is a second exemplary embodiment of theinvention for an RF feedback circuit 600 are correctable. As shown inFIG. 6, an RF signal to be amplified, RF_(in), is applied to the inputport 600A of an RF feedback circuit 600, and is electrically connectedto the input splitter 610. The primary output port 610A of the inputsplitter 610 is electrically connected to first port 620A of thesummation circuit 620. The secondary output port 610B of the inputsplitter 610 is connected to the first port 644A of the delta generatorcircuit 644, via signal attenuator 674.

The output port of the summation circuit 620 is connected to the inputport of the power amplifier 630 via the first phase shifter circuit 625.The RF input signal, RF_(in), is amplified by the power amplifier 630,and electrically coupled to the output port 600C of the RF feedbackcircuit 600 via the output tap coupler 615. The second output port ofthe output tap coupler 615 represents a predetermined portion of theoutput signal from the power amplifier 630 and is therefore anattenuated replica of the output signal, RF_(out), at the output port600C. The output signal being RF_(out)=

(RF_(in))+∂, where

( ) represents the amplification process and ∂ represents distortionadded by the power amplifier 630.

The second output port of the output tap coupler 615 is then coupled toan attenuator 640 and electrically coupled to the second port 644B ofthe delta generator circuit 644. The attenuator 640 thereby furtherreducing the power of the replica of the output signal, RF_(out), toapproximately that of the tapped portion of the RF signal, RF_(in).Accordingly, the delta generator circuit 644 produces an output signalat output port 644C which essentially represents distortion generatedwithin the power amplifier 630 as the input signal, RF_(in), issubtracted from the tapped and attenuated replica of the output signal,RF_(out).

This output signal ∂ ∝

RF_(out)−RF_(in)

is then electrically coupled to the feedback processing circuit 6100.The output signal ∂ is first coupled to the vector demodulator 660 whichconverts this output signal ∂ to I and Q baseband error signals ∂_(I)and ∂_(Q), which are output signals from I error port 660A and Q errorport 660B respectively. The I baseband error signal ∂_(I) iselectrically coupled via first baseband amplifier 672 before beingfiltered by first baseband filter 676 and coupled to the I input port690A of the vector modulator 690. Similarly, the Q baseband error signal∂_(Q) is electrically coupled via second baseband amplifier 670 beforebeing filtered by second baseband filter 674 and coupled to the Q inputport 690B of the vector modulator 690. The output signal from the vectormodulator 690 is boosted by the error amplifier 695 before beingelectrically coupled to the first port 620A of the summation circuit620. This subtracts the error signal ∂ generated by the power amplifier630 from the RF signal to be amplified RF_(in) to provide correction foroperation of the power amplifier 630.

The local oscillator signal required by the vector demodulator 660 andvector modulator 690 is provided by the local oscillator 685. Whilst thelocal oscillator 685 is shown directly coupled to the vector modulator690, it is electrically connected to the vector demodulator 660 via asecond phase shifter circuit 680, which ensures the appropriate phase ofthe local oscillator signal to demodulate the I and Q signals ∂_(I) and∂_(Q) respectively.

Fulfilling high loop gain, for high distortion suppression, whilstavoiding oscillation is challenging, as will be evident to one skilledin the art, because the amplifier may have several stages, and narrowband matching, so that the phase through the amplifier changes quicklywith frequency, in other words, the amplifier may have significantdelay. As such, stability of the feedback loop is provided by the firstand second baseband filters 674 and 676, these reducing the overall gainof the feedback loop to below unity for frequencies wherein the phaseshift around the feedback loop from the output port of the poweramplifier 530 through to its input port is approximately zero, therebysuppressing the oscillation of the feedback loop. However, unlike priorart solutions these first and second baseband filters 674 and 676,respectively, are only filtering the baseband error signal generatedfrom the delta generator circuit 644 and converted to baseband by thevector demodulator 660. Beneficially, the critical design aspect of thisfilter is transformed from “flatness of frequency response” to theprovisioning of sharp cut-off allowing high in-band loop gain but rapidreduction in out-of-band gain as required to meet loop stabilityrequirements.

Whilst the exemplary embodiment of the invention has been presented inrespect of FIG. 6 for the Cartesian error feedback method, alternativelyother coordinate systems for the provision of baseband signals areemployed. As such, the error feedback approach of the invention isapplicable to polar coordinate modulation formats, and optionallyarbitrary coordinate modulation formats. Further, as with the RFfeedback circuit 500 of FIG. 5 the RF feedback circuit 600 can alsosupport WiMAX subscriber units having over 50 dB of dynamic range inoutput power, the PA within such units will typically operate at lowoutput power, wherein the main power amplifier is typically relativelylinear, to high output power, wherein it is typically highly non-linear.As such, the RF feedback circuit 600 similarly optionally supports thetwo modes of operation presented supra in respect of the RF feedbackcircuit 500.

Further, the RF feedback circuit 600 supports localized control from thecontroller circuit 6000, as well as control from other elements of thedevice within which the RF feedback circuit 600 is operating. Thecontroller circuit 6000 as shown receives input control signals at firstand second control input ports 6000A and 6000B, respectively. The firstcontrol input port 6000A is electrically connected to first monitoringpoint 6100A, which is placed after the first baseband filter 676 on theI path of the feedback loop. As such, the first control input port 6000Areceives an error signal I_(err), where I_(err) ∝

∂ {circle around (×)} f(LO)|φ=

, and represents the in-phase portion of the baseband converted errorsignal ∂. Similarly, the second control input port 6000B is electricallyconnected to second monitoring point 6100B, which is placed after thesecond baseband filter 674 on the Q path of the feedback loop. As such,the second control input port 6000B receives an error signal Q_(err),where Q_(err) ∝

∂ {circle around (×)} f(LO)|φ=90

, and represents the quadrature portion of the baseband converted errorsignal ∂. Alternatively, the first and second monitoring points 6100Aand 6100B may be placed before the first and second baseband filters 674and 676.

Based upon these error signals I_(err) and Q_(err) the controllercircuit 6000 determines the appropriate settings for control signalsprovided to several elements of the RF feedback circuit 600. A first setof control signals are provided from the controller outputs 6000C, 6000Dand 6000E, the first controller output 6000C providing the controlsetting for the signal attenuator 635, the second controller output6000D providing control of phase shifter 625 and the third controlleroutput 6000E providing the control setting for the attenuator 640. Thecontroller monitors the signals I_(err) and Q_(err) at its inputs6000A/B and adjusts the three elements 674, 625 and 640 in order tominimize the power of (I_(err) ²+Q_(err) ²). In this manner thecontroller circuit 6000 balances in both amplitude and phase the tappedportions of the applied RF signal RF_(in), and output of the RF feedbackcircuit 600 being RF_(out) fed to the delta generator circuit 644. Theoutput signal 644C of the delta generator circuit 644 then contains aminimum component of the wanted signal, and consists predominantly ofthe error signal ∂.

The controller circuit 6000 also provides a control signal fromcontroller output 6000F which provides the control setting for thesecond phase shifter circuit 680. Accordingly the controller circuit6000 is able to control the phase shift of the local oscillator signalprovided to the vector demodulator 660 and thereby adjusts the phaseshift of the combined signal applied to the input of the power amplifier630, RF_(applied)=RF_(in)+∂ such that the feedback applied is in thecorrect phasing to improve the quality of the output RF_(out) of the RFfeedback circuit 600. The algorithm used to adjust 660 may be based onfactory calibration, optionally using other known pieces of a-prioriinformation such as, but not limited to, operating frequency; or by someother means.

The second exemplary embodiment of invention presented in respect of theRF feedback circuit 600 provides control of the first and second phaseshifters 625 and 680, respectively, along with control of theattenuation settings of the signal attenuator 635 and attenuator 640. Inorder to minimize the distortion in the RF output signal 600C arisingfrom the power amplifier 630, the controller circuit 6000 seeks toadjust the first phase shifter 625, signal attenuator 635, andattenuator 640 to null the signals detected at first and secondmonitoring points 6100A and 6100B. The controller circuit 6000 adjustssecond phase shifter 680 for both increasing loop stability and toapproximately optimize error cancellation.

Optionally, the controller circuit 6000 provides only control of some ofthese circuit elements, or alternatively may provide additional controlsuch as providing control of the phase of the local oscillator 680applied to the vector modulator 630, and an attenuator provided betweenthe summation circuit 620 and power amplifier 630 (not shown forclarity). Additionally, the controller optionally establishes updates todifferent control loops at different rates, such as, for example,providing frequent updates of the phase of the local oscillator 680applied to the vector demodulator 660 and infrequent updates to thesignal attenuator 635 and attenuator 640.

Optionally, the controller circuit 6000 receives additional inputcontrol signals such as a transmit enable, therein providing thecontroller circuit 6000 with information as to status of the transmitterto which the RF feedback circuit 600 is connected. This informationallows timing for controlling adjustment triggers, implementingadjustments of power level, or enabling the provisioning of a dedicatedtest signal such as outlined supra to be established by the controllercircuit 6000. Alternatively, the controller circuit 6000 optionallyreceives a power setting signal indicating the intended target outputpower of the RF signal provided at the output port 600C, RF_(out)

Numerous other embodiments may be envisaged without departing from thespirit or scope of the invention.

1. A method comprising: amplifying an applied RF input signal togenerate an RF output signal with an amplifier; generating an errorsignal by combining a predetermined portion of the RF output signal witha predetermined portion of a supplied RF input signal; modifying theerror signal to generate a modified error signal, the modified errorsignal generated by down-converting the error signal, processing thedown-converted error signal and up-converting the processeddown-converted error signal; reducing distortion in the RF output signalby combining the modified error signal with the supplied RF input signalto provide the applied RF input signal; and controlling an aspect of atleast one of the generation of the modified error signal, generation ofthe error signal, and the applied RF input signal at least in dependenceupon a magnitude of the down-converted error signal.
 2. A methodaccording to claim 1 wherein, processing the down-converted error signalcomprises modifying at least one of a time delay, an amplitude, and aphase of at least a predetermined baseband portion of the down-convertederror signal.
 3. A method according to claim 1 wherein, controlling anaspect of the at least one of the generation of the modified errorsignal, generation of the error signal, and the applied RF input signalat least in dependence upon a magnitude of the down-converted errorsignal comprises adjusting the phase of an oscillator providing anoscillator signal employed in at least one of down-converting the errorsignal and up-converting the processed down-converted error signal.
 4. Amethod according to claim 1 wherein, controlling an aspect of the atleast one of the generation of the modified error signal, generation ofthe error signal, and the applied RF input signal at least in dependenceupon a magnitude of the down-converted error signal comprises modifyingat least one of a time delay, an amplitude, and a phase of thepredetermined portion of the supplied RF input signal and thepredetermined portion of the RF output signal.
 5. A method according toclaim 1 wherein, processing the down-converted error signal comprisesfiltering the down-converted error signal.
 6. A method according toclaim 5 wherein, filtering the baseband error signal comprises filteringwith a low pass filter, the low pass filter characterized by at least acutoff frequency, wherein the cutoff frequency multiplied by the timedelay from an output port of the amplifier to an input port of theamplifier is less than is 25 MHz.ns.
 7. A method according to claim 1wherein, controlling comprises controlling irrespective of the contentof the applied RF signal.
 8. A method according to claim 1 wherein,controlling comprises controlling when the applied RF signal comprisesdata to be transmitted.
 9. A circuit comprising: an amplifier foramplifying an applied RF input signal to generate an RF output signal;an error signal circuit for generating an error signal by combining apredetermined portion of the RF output signal with a predeterminedportion of a supplied RF input signal; an error signal modificationcircuit for modifying the error signal to generate a modified errorsignal, the modified error signal generated by down-converting the errorsignal, processing the down-converted error signal and up-converting theprocessed down-converted error signal; an error combiner circuit forreducing distortion in the RF output signal by combining the modifiederror signal with the supplied RF input signal to provide the applied RFinput signal; and a controller for controlling an aspect of at least oneof the generation of the modified error signal, generation of the errorsignal, and the applied RF input signal at least in dependence upon amagnitude of the down-converted error signal.
 10. A method comprising;(a) amplifying a microwave signal with at least an amplifier to providean output signal, the output signal being substantially an amplifiedreplica of the microwave signal; (b) providing an input signal, theinput signal forming a portion of the microwave signal to be amplified;(c) extracting a predetermined portion of the output signal; (d)combining a predetermined portion of the input signal with thepredetermined portion of the output signal to form an error signal, theerror signal substantially suppressing the input signal component andsubstantially unaffecting distortion components of the output signal;(e) down-converting the error signal according to a predeterminedcoordinate system to provide down-converted error components; (f)processing the down-converted error components to provide modifieddown-converted error components; (g) up-converting the modifieddown-converted error components according to the predeterminedcoordinate system to provide a modified error signal; (h) combining themodified error signal with the input signal to form the microwave signaland provide a reduction in the distortion components of the outputsignal; and (i) providing at least a control signal, the control signalbeing one of a plurality of control signals, each control signal forcontrolling an aspect of at least one of steps (a) through (h) and beingdetermined in dependence upon a measure of one of the down-convertederror components and modified down-converted error components.
 11. Amethod according to claim 10 wherein, controlling an aspect of at leastone of steps (a) through (h) comprises controlling an aspect of at leastone of amplifying the microwave signal, generating the predeterminedportion of the input signal, extracting the predetermined portion of theoutput signal, down-converting the error signal, up-converting themodified down-converted error components and processing thedown-converted error components.
 12. A method according to claim 10wherein, step (h) comprises determining the control signal irrespectiveof the content of the input signal.
 13. A method according to claim 10wherein, step (h) comprises determining the control signal when theinput signal represents live traffic.
 14. A method according to claim 10wherein, step (h) further comprises determining the control signal independence upon a first mode of operation of the amplifier whenprocessing live traffic and a second mode of operation when other thanprocessing live traffic.
 15. A method according to claim 14 wherein,determining the control signal in the second mode operation comprises atleast one of maintaining the control signal at a value established fromthe first mode of operation and providing a predetermined test signal asthe input signal and determining the control signal by nulling amagnitude of the down-converted error components.
 16. A method accordingto claim 10 comprising; performing steps (e) through (g) in a first modeof operation; and other than performing steps (e) through (g) in asecond mode of operation.
 17. A method according to claim 16 wherein,other than performing steps (e) through (g) comprises at least one ofdeactivating a predetermined portion of a circuit implementing themethod and reducing a supply voltage to a predetermined portion of acircuit implementing the method to a predetermined value.
 18. A methodaccording to claim 16 wherein, determining the mode of operationcomprises determining the mode of operation in dependence of at leastone of a target output power of the amplifier and a target powerconsumption of a circuit of which the amplifier forms part thereof. 19.A method according to claim 10 wherein, providing at least a controlsignal comprises providing a control signal for controlling the phase ofan oscillator signal for use in at least one of step (e) and step (g).20. A method according to claim 10 wherein, providing at least a controlsignal comprises providing a control signal for controlling at least oneof the power and phase of the microwave signal, the power ofpredetermined portion of the input signal, and power of thepredetermined portion of the output signal.
 21. A method according toclaim 10 wherein, providing at least a control signal comprisesproviding a first control signal for controlling the power of themicrowave signal and a second control signal for controlling the powerof the predetermined portion of the output signal, the first and secondcontrol signals for maintaining the power of the microwave signal andthe power of the predetermined portion of the output signal according toa predetermined relationship.
 22. A method according to claim 10comprising; reducing distortion in the output signal by repeating step(i) and determining for each step (i) the at least a control signal soas to minimize the measure of the down-converted error components.
 23. Amethod according to claim 10 comprising, increasing linearity of theamplification by repeating step (i) and determining for each step (i)the at least a control signal so as to minimize the measure of thedown-converted error components.
 24. A method according to claim 10wherein, down-converting the error signal according to a coordinatesystem comprises down-converting according to standard modulationformat, the modulation format being at least one of quadraturemodulation, polar modulation, and modulation in a predeterminedcoordinate system.
 25. A method according to claim 10 wherein, providingat least a control signal comprises providing a first control signal forcontrolling the power of the predetermined portion of the input signaland a second control signal for controlling the power of thepredetermined portion of the output signal, the first and second controlsignals for increasing suppression of the input signal component andincreasing the power of the distortion components in step (d).
 26. Amethod according to claim 10 wherein, step (h) comprises processing themodified error signal prior to combining it with the input signal.
 27. Amethod according to claim 10 wherein, processing a down-converted errorcomponent comprises at least one of adjusting the phase of thedown-converted error component, adjusting the amplitude of thedown-converted error component, and filtering the down-converted errorcomponent.
 28. A method according to claim 27 wherein, filtering thedown-converted error component comprises filtering the down-convertederror component with a low pass filter, the low pass filtercharacterized by at least a cutoff frequency, the product of the cutofffrequency, in MHz, multiplied by the time delay, in ns, of steps (d)through (g) is less than 25 MHz.ns
 29. A method according to claim 10wherein, step (h) comprises delaying the input signal prior to combiningit with the modified error signal, the delay being approximately that ofsteps (a) and steps (d) through (g).
 30. A method according to claim 10wherein, executing steps (a) through (i) is by using at least asemiconductor integrated circuit.
 31. A method according to claim 10wherein, executing steps (a) through (i) comprises; providing anelectrical circuit; providing a first predetermined portion of theelectrical circuit as portion of a first integrated circuit; providing asecond predetermined portion of the electrical circuit as a portion of asecond integrated circuit; and assembling the first and secondintegrated circuits, the assembly minimizing the time delay from theamplifier output port to the amplifier input port in executing steps (a)and steps (c) through (h).
 32. A method according to claim 10 wherein,executing steps (a) through (h) comprises; providing an electricalcircuit; providing a first predetermined portion of the electricalcircuit as a portion of a first circuit package; providing a secondpredetermined portion of the electrical circuit as a portion of a secondcircuit package; and assembling the first and second circuit packages,the assembly reducing the time delay from the amplifier output port tothe amplifier input port in executing steps (a) and steps (c) through(h).
 33. A method according to claim 32 comprising; providing apredetermined portion of the electrical circuit as a portion of itsrespective circuit package comprises placing the respective portion ofthe electrical circuit within the circuit package to reduce the timedelay associated with the portion of the electrical circuit and itsrespective circuit package.